Process and device for converting digitally modulate high-frequency reception signals

ABSTRACT

In a receiver for digitally modulated signals in mobile communication systems, the reception signal can be represented as a complex vector. The signal undergoes homodyne or heterodyne incoherent conversion into the baseband by means of a mixing stage with mixers M. This involves splitting into the real component and the imaginary component. The signal components are filtered by low-pass filters (TP) and digitized with analog-digital converters (AD). The sampled values are converted into the magnitude and angle of the vector. The magnitude value (B) controls the gain of the preamplifier (VV) and of the low-pass filters (TP), the reception data (D) are recovered from the difference between two successive angle values (W).

BACKGROUND OF THE INVENTION

The invention relates to a method of converting a digitally modulatedreception signal from the radio-frequency range, in which the receptionsignal, which can be represented as a complex vector, is convertedin-coherently into the baseband. The invention relates furthermore to areceiver for digitally modulated signals in a mobile communicationsystem, the antenna of which is connected to at least one mixing stagefor the reception signal which can be represented as a complex vector,the mixing stage being followed by two low-pass filters which areconnected on the output side via an analog-digital converter in eachcase to a value converter.

A receiver operating by such a method is known from PCT InternationalPublication WO 86/3356. The reception signal is converted by two mixerswith a local oscillator in correct phase relation and phase-shifted by90°. Low-pass filters and analog-digital converters are followed by adigital filter. The decoding of the transmitted bits is performed with aROM, operating as a value converter, which is followed by a multiplexer(FIG. 17).

A reception method and a receiver for cordless telephones is describedin Telcom Report 10 (1987), issue 2, pages 130 to 137. The receptionsignal is mixed into the baseband via a radio-frequency stage and aplurality of intermediate-frequency stages. The demodulation isperformed in phase-locked loop.

In general, digitally modulated signals are processed by synchronous or,more rarely, asynchronous heterodyne receivers. An image frequencysuppression is performed at the radio-frequency stage, the mainselection at one to two intermediate-frequency stages. Limiteramplifiers or gain controls provide the necessary compensation of thedynamic range of the input signal. Linear modulation methods divide intotwo-stage methods with binary amplitude modulation (BAM) and binaryphase-shift keying (BPSK), four-stage methods with quadrature amplitudemodulation (QAM) and quadrature phase-shift keying (QPSK), as well asmultistage modifications. Customary non-linear modulation methods areFSK or frequency modulation-like methods, such as minimum-shift keying(MSK) and Gaussian minimum-shift keying (GMSK). The demodulation isperformed for non-linear modulation methods by means of phase lock loop,Costa's loop or discriminators. For linear modulation methods,synchronous quadrature demodulators are used.

SUMMARY OF THE INVENTION

The invention is based on the object of permitting the processing ofsignals digitally modulated at two or more stages with a receiver.

According to the present invention, this is achieved by arranging thatthe reception signal is split into a real component and an imaginarycomponent, continuous in time and amplitude, the phase difference perbit period due to the incoherent conversion being kept to less than halfthe phase shift caused by the modulation per bit, the signal componentsare subsequently amplified, filtered and sampled, the sampling timebeing determined within a preceding synchronization time, the sampledsignals are subsequently digitized and the sampled values arevalue-converted as a pair of binary numbers for the magnitude and angleof the vector, the magnitude value serving for gain control andreception data being recovered from the difference between twosuccessive angle values after classification.

Due to a homodyne mode of operation, the occurrence of an imagefrequency is avoided. The incoherent demodulation avoids complex carriersynchronization and allows DC voltage-free signal paths. The phasedifference is kept small by frequency correction or oscillators withexact frequency stability. In spite of using only one mixing stage, withthis method of reception virtually all two-stage and multistage, linearand non-linear modulation methods can be employed.

An efficient angle decoding beyond 360° is achieved by an angle valuebeing buffer-stored over a bit period, subtracted from the followingvalue by two's complement and the reception data recovered from aclassification of the result dependent on the modulation method used.The adaptation to the individual modulation method is performed by meansof a programmable setting of the classification, as a result of which nomodification of the circuitry is necessary.

It is advantageous that the sampling time for the angle value of alinearly modulated reception signal is determined within asynchronization time before the transmission of valid data by thevariation of the magnitude of the reception signal being modulated witha synchronization sequence and, after conversion into the baseband, thesign function formed and digitally differentiated, a zero pulse sequencebeing formed, the timing of the pulses in relation to the system clockmeasured and averaged over several bits. From the rotation of the vectorof the reception signal, a correction of the phase difference is derivedby incoherent conversion.

Furthermore, it is advantageous that the sampling time for the anglevalue of a non-linear modulated reception signal is determined within asynchronization time before the transmission of valid data by thecarrier which has been angle-modulated with the synchronization sequencebeing oversampled after conversion into the baseband and from it a phasedeviation obtained, from this a phase difference being formed bysubtracting over a bit period, the angle sign function of which phasedifference is compared in terms of phase with the data clock and thephase shift averaged over several bits.

The object of the invention is also achieved by the low-pass filtersbeing provided with variable gain, by the value converter includingmeans for determining the magnitude and angle of the vector of thereception signal from its real component and its imaginary component andbeing connected to a decoder, which contains a classifier, for receptiondata recovery, and by an adjusting device being connected to the valueconverter and the decoder for determining the sampling time.

The conversion into the baseband using a mixing stage dispenses withradio-frequency circuitry. After the radio-frequency prestage, thecircuit components can be fully integrated by Si-bipolar or MOStechnology. It is no longer required to tune radio-frequency orinter-mediate-frequency stages. The control behavior of the digitalamplitude control is programmable and can be changed withoutmodification of the circuitory. Undesired signal components are removedby the low-pass filters.

It is advantageous that the mixing stage is designed as a quadraturedemodulator, which comprises two mixers which are connected to a localoscillator for feeding with an oscillator signal phase-shifted by 0° and90° and the local oscillator has a tuning input which is connected to anoffset correction device. The local oscillator is not coupled inlocked-phase relation to the input signal, thus dispensing withcircuitry for synchronization. The carrier frequency offset from thephase shift of the incoherence of the frequency of input signal andlocal oscillator is minimized by means of the tuning input. Its value isobtained when determining the sampling time and no longer has to beadditionally calculated.

It is advantageous that the gain factor of the low-pass filters can beset digitally and the value converter is coupled back to the low-passfilters for this control. To eliminate DC voltage components, thelow-pass filters contain DC isolating capacitors. They therefore act asbandpass filters with a band of several decades.

To reduce the current consumption and the processing of high data rates,it is advantageous that the value converter comprises two permanentmemories, the data addresses of which are formed by the numerical valuesfor the real component and the imaginary component of the sampledvalues, and that the magnitude value and the angle value are availableat the data output of the respective permanent memory.

In order to be able to extend the demodulation of the reception signalto extremely high frequencies, it is advantageous that the mixing stageis preceded by at least one intermediate-frequency converter. The rangeof application of the receiver is extended by the fact that it can beadapted by different intermediate-frequency converters to differentreception frequencies.

BRIEF DESCRIPTION OF THE DRAWINGS

The features of the present invention which are believed to be novel,are set forth with particularity in the appended claims. The invention,together with further objects and advantages, may best be understood byreference to the following description taken in conjunction with theaccompanying drawings, in the several Figures in which like referencenumerals identify like elements, and in which:

FIG. 1 shows a block diagram of the first illustrative embodiment,

FIG. 2 shows the timing structure of the reception signal,

FIG. 3 shows the phase response in angle modulation for determining thesampling time and

FIG. 4 shows the signal response for determining the sampling time inthe case of linear modulation methods,

FIG. 5 shows a block diagram of the second illustrative embodiment.

DESCRIPTION OF THE PREFERRED EMBODIMENT

The block diagram of a receiver for signals digitally modulated at twoand four stages for use in mobile communication systems is shown inFIG. 1. The first illustrative embodiment is explained with reference tothis figure with incidental explanation of FIGS. 2-4 and with tables. Itis a homodyne, incoherent, digital vector receiver with a mixing stage.Analog signal paths are marked as single lines, digital data paths aremarked as double lines. The control branches are correspondingly dashed.

A preamplifier VV, connected to an antenna A, boosts the input levelwith low noise and low intermodulation and at the same time effects adecoupling of the antenna A from the inputs of the mixing stage. Thisradio-frequency prestage includes only a fixed, rough preselection ofthe input frequency range. In addition, the preamplifier VV is providedwith a variable gain in order to permit a partial compensation of thedynamic range of the input signal.

In the mixing stage, the input signal is converted in one step from thereceive frequency f_(E), about 2 GHz, into a baseband. In order topreserve the full informational content, the reception signal must besplit into a real component and an imaginary component. This takes placeby means of a quadrature demodulator, which comprises a power divider LTand two identical mixers M, which are fed with a signal phase-shifted by0° and a signal phase-shifted by 90° of a local oscillator VCO with anoscillator frequency f_(LO).

The local oscillator VCO is not coupled in locked-phase andlocked-frequency relation to the input carrier frequency. In order tokeep a carrier frequency offset between the receive frequency f_(E) andthe oscillator frequency f_(LO) as small as possible, a tuning input isprovided for a rough frequency correction at the local oscillator VCO.The tuning voltage is determined in an offset correction device OKE.However, the frequency difference between receive frequency f_(E) andoscillator frequency f_(LO) should not be more than 5 ppm.

After the mixers M, the useful signal, heterodyned with noise andadjacent-channel signals, is available in the baseband. Two low-passfilters TP with linear delay effect the main selection of the receiver.They contain DC isolating capacitors for suppressing DC components ofthe useful signal and therefore act as broadband band pass filters. Thegain of the individual filter stages can be set digitally, so that heretoo a dynamic-range compensation can be carried out. After the filterstages, the real component and the imaginary component of the receptionsignal would be available in band-limited, amplitude-continuous andtime-continuous form.

After the low-pass filters TP, the signals are periodically sampled witha system clock ST in the real branch and the imaginary branch bysample-and-hold sections SH and are digitized in two analog-digitalconverters AD. Due to the dynamic-range compensation at the preamplifierVV and at the low-pass filters TP, the dynamic-range fluctuations arerestricted to a few dB and a resolution of the analog-digital convertersAD of 5-8 bits is adequate. The system clock ST is generated by a clockgenerator TG and has a frequency of 8.8 MHz. Thereafter, the realcomponent and the imaginary component of the input signal are availableat any time as binary numerical values. The data rate is 1.1 Mbit/secwith a data clock DT of 1.1 MHz, whereby an 8-fold oversampling isaccomplished by the system clock ST.

Since the information received is coded as the difference in anglebetween sampling times, a value conversion table in permanent memoriesROM is used to convert the real component and the imaginary componentinto the magnitude and angle of a complex vector. The numerical valuesfor the real component and the imaginary component form the dataaddresses of the permanent memories ROM; magnitude value B and anglevalue W are obtained at the data output. The magnitude value B is usedfor setting the gain in the preamplifier VV and in the low-pass filtersTP of the main selection. The angle information is coded in such a waythat N bits with a range of values of 2^(N) correspond to the full angleof 360° (see Table 1). This coding allows correct-sign, binary angleaddition and subtraction by two's complement beyond 360°.

    ______________________________________                                        Angle            Binary code                                                  ______________________________________                                        0°        0000                                                         +22.5°    0001                                                         +90°      0100                                                         +180° = -180°                                                                    1000                                                         +270° = -90°                                                                     1100                                                         +337.5° = -22.5°                                                                 1110                                                         +360° = 0°                                                                       0000                                                         ______________________________________                                         Tab. 1: Angle coding with 4 bits (excerpt)                               

A subtracter SB forms the difference between the current angle value Wand the angle value W of the previous sampling time, stored in a buffermemory ZS. A classification, dependent on the selected modulationmethod, of the magnitude of the difference in angle Δφ in decision areasE (Table 2) then supplies demodulated reception data D, the samplingtime of which is determined in an adjusting device JE. For this purpose,a classifier K and a latch L are combined with the subtracter SB to forma decoder DC. The value is selected from the latch L. On account of thedecision areas E, the vector receiver can be changed over automaticallyby a following microcomputer to different modulation methods. This makesit possible to set up a multi-standard device.

    ______________________________________                                        Modulation method                                                                         Decision areas E                                                  ______________________________________                                        BAM, BPSK   Δφ ≦ +90°, Δφ > +90°                 4                                                                 QAM, WPSK   -45° ≦ Δφ < +45°, +45°                  ≦ Δφ < +135°                                          +135°≦ Δφ < +225°,                             -135° ≦ Δφ < -45°                  MSK, GMSK   0° ≦ Δφ < +180°, -180°                  ≦ Δφ < 0°                                 ______________________________________                                         Tab. 2: Decision area of the vector receiver for twostage and fourstage       modulation methods                                                       

In order to make a correct assignment by the classifier K possible, thephase shift which is produced by the incoherence of input signal andlocal oscillator VCO per bit must be less than half the angle of adecision area E. It follows from this that the maximum permissibledifference in frequency f₀,max between reception signal and localoscillator signal is ##EQU1##

In the equation, Δφ_(E) denotes the angle of a decision area E andT_(bit) denotes the bit period, which is the reciprocal of the dataclock DT.

On account of the difference in frequency between input frequency f_(E)and oscillator frequency f_(LO), the differences in angle

    Δφ=Δφ.sub.bit +2·π·(f.sub.E -f.sub.LO)·t

for modulation with positive and negative phase shift Δφ_(bit) are notsymmetrical. The carrier frequency offset Δf is calculated by the offsetcorrection device OKE from the difference in phase deviations as##EQU2## With known voltage dependence of the frequency of the localoscillator VCO, a frequency correction signal is derived as tuningvoltage.

As FIG. 2 shows, a synchronization sequence has to be transmitted at thebeginning of each reception sequence ES to determine the sampling time.This is initiated by gating out the transmission signal for a shorttime. This produces the timing structure of the reception signal withgating-out time AZ, synchronization time SZ and bit-transmission timeBZ. Over which time the sampling time is then valid is decided by thestability of the transmission data clock and the reception data clock,as well as the fineness of the time resolution for the sampling within abit. The number N of the validly received bits is obtained as ##EQU3##Δf_(clock),max being the maximum difference in clock frequency betweentransmitter and receiver and OV (oversampling) being the number ofpossible sampling times per bit. After a maximum of N data bits, a freshsynchronization sequence must be transmitted in order that the receivercan update the sampling time. With OV=8 and a data clock accurate towithin 50 ppm (Δf_(clock),max =50·10⁻⁶) and T_(bit) =909·1·10⁻⁹ secthere are 1250 validly received bits (N).

For the linear and non-linear modulation methods, varioussynchronization methods are used for determining the sampling time. Inthe case of the non-linear modulation methods (MSK, GMSK), as shown inFIG. 3 for MSK, a periodic phase deviation PH can be produced bymodulation with a synchronization sequence SF corresponding to a datasequence d(t)=+1,-1,+1,-1,+1 . . . . In the case of non-linearmodulation methods, the magnitude of the vector is constant (constantenvelope). In the demodulation, the phase deviation PH increasesconstantly as a consequence of the carrier frequency offset. By 8-foldoversampling and phase subtraction with the value preceding by 1 bit foreach sampling time within a bit, a phase difference PD can be formed.The angle sign function WVF thereof corresponds to the synchronizationsequence SF shifted by T_(bit) /2. By phase comparison with the dataclock DT and forming the average over time, the sampling time can bedetermined very accurately therefrom, even in the case of noisy inputconditions.

In the case of the linear modulation methods (BAM,BPSK,QAM,QPSK), adata-clock periodic zero transition of the reception signal can beproduced by modulation with the synchronization sequence SF, as shown inFIG. 4 for linear amplitude modulation with suppressed carrier. Thevariation of the carrier frequency offset Δf, as well as the variationof the real component RT and the imaginary component IT of the receptionsignal are shown. The magnitude of the vector is variable in the case oflinear modulation methods. After forming the sign function VZF of thereal component RT and of the imaginary component IT, narrow pulses aregenerated by means of digital differentiation at the time of the zerotransition. They form a zero transition pulse sequence NF. By measuringthe position in time of the pulses in relation to the system clock andaveraging over a number of bits, the sampling time is obtained with highreliability, even in the case of noisy reception conditions.

The automatic gain control is a controller R of a digital type, shown inFIG. 1. It has the task of keeping the signal level, which fluctuatesduring operation, constant to within a few dB when driving thedigital-analog converters DA in a medium range. As actual level value,the magnitude value B, formed in the permanent memory ROM, is read intothe controller R. The controller R must operate so slowly that it cannotfollow the brief dips in level before the synchronization times. Toreduce intermodulation interferences, the non-band-limited signal poweravailable after the mixers M is used as the parameter for dividing thegain over the RF prestage and the main selection stages. The parameteris determined by forming the magnitude of the reception vector in ascaling device NV.

FIG. 5 shows the diagrammatic block diagram of a heterodyne vectorreceiver as second illustrative embodiment. It operates like the vectorreceiver of the first illustrative embodiment with incoherentdemodulation. The signal received via the antenna A is reduced by meansof an intermediate-frequency converter ZFU from 60 GHz to 2 GHz.Consequently, a vector receiver of the same design as in the firstillustrative embodiment can be used modularly for the reception ofsatellite signals. The processing of the reception signal after thepreamplifier VV remains the same. The intermediate-frequency converterZFU contains an input amplifier EV, which is followed by a premixer VM.The premixer VM is connected to an intermediate-frequency oscillatorZFO.

The invention is not limited to the particular details of the method andapparatus depicted and other modifications and applications arecontemplated. Certain other changes may be made in the above describedmethod and apparatus without departing from the true spirit and scope ofthe invention herein involved. It is intended, therefore, that thesubject matter in the above depiction shall be interpreted asillustrative and not in a limiting sense.

What is claimed is:
 1. A method of converting a digitally modulatedreception signal from a radio-frequency range, comprising the steps of:converting the reception signal, which can be represented as a complexvector, incoherently into a baseband, the reception signal being splitinto a real component and an imaginary component, continuous in time andamplitude, a phase difference per bit period due to the incoherentconversion being kept to less than half a phase shift caused by amodulation per bit, the real and imaginary components being subsequentlyamplified, filtered and sampled, a sampling time being determined withina preceding synchronization time, the sampled real and imaginarycomponents being subsequently digitized thereby providing sampled valuesthat are value-converted as a pair of binary numbers for magnitude valueand angle value of the vector, using the magnitude value to adjust alevel of the real and imaginary components in the step of amplifying,filtering and sampling the real and imaginary components, the magnitudevalue serving for gain control thereof; and recovering reception datafrom a difference between two successive angle values of respectivelytwo successive sample times after a classification, dependent on aselected modulation method, of the magnitude of the difference betweensaid two successive angle values.
 2. The method as claimed in claim 1,wherein the angle value is buffer-stored over a bit period, subtractedfrom a following angle value by two's complement and the reception datarecovered from the classification of the result of the subtractiondependent on the modulation method used.
 3. The method as claimed inclaim 1, wherein the sampling time for the angle value of a linearlymodulated reception signal is determined within a synchronization timebefore transmission of valid data by a variation of the magnitude of thereception signal being modulated with a synchronization sequence and,after conversion into the baseband, a sign function formed and digitallydifferentiated, a zero pulse sequence being formed, timing of the pulsesof the zero pulse sequence measured in relation to a system clock andaveraged over several bits.
 4. The method as claimed in claim 1, whereinthe sampling time for the angle value of a non-linearly modulatedreception signal is determined within a synchronization time beforetransmission of valid data by a carrier which has been angle-modulatedwith a synchronization sequence being oversampled after conversion intothe baseband from which a phase deviation is obtained, from the phasedeviation a phase difference being formed by subtracting over a bitperiod, an angle sign function of the phase difference being compared interms of phase with a data clock and the phase shift averaged overseveral bits.
 5. A receiver for receiving digitally modulated signals ina mobile communication system, comprising: an antenna connected to atleast one mixing stage for a reception signal, which can be representedas a complex vector, the mixing stage having two outputs connectedrespectively to two low-pass filters, each of which are connected on anoutput side via an analog/digital converter to a value converter,wherein the low-pass filters are provided with variable gain, whereinthe value converter includes means for determining a magnitude value andan angle value of the vector of the reception signal from its realcomponent and an imaginary component of the reception signal and isconnected to a decoder for reception data recovery, and wherein anadjusting device is connected to the value converter and the decoder fordetermining a sampling time that is supplied to the decoder wherein again factor of the low-pass filters is digitally set and the valueconverter is coupled back to the low-pass filters for this control, andwherein the low-pass filters contain DC isolating capacitors.
 6. Thereceiver as claimed in claim 5, wherein the mixing stage is a quadraturedemodulator having two mixers which are connected to a local oscillatorfor feeding with an oscillator signal phase-shifted by 0° and 90°,respectively and wherein the local oscillator has a tuning input, whichis connected to an offset correction device.
 7. The receiver as claimedin claim 5, wherein the value converter has two permanent memories, dataaddresses of which are formed by numerical values of sampled values forthe real component and the imaginary component and wherein the magnitudevalue and the angle value are available at data outputs of therespective permanent memories.
 8. The receiver as claimed in claim 5,wherein the mixing stage is preceded by at least oneintermediate-frequency converter.